Voltage level shifter for arbitrary input signals

ABSTRACT

Methods, systems, and devices are described for providing voltage level shifting that may operate reliably and at low power, even at high voltages and/or high switching frequencies. Embodiments receive an input signal representing input information, and effectively generate two voltage responses as a function of the input signal. Each voltage response includes exponential terms as a function of resistive and capacitive loading effects of components of the embodiments. A combined response signal is generated substantially as a superposition of the first response signal and the second response signal. A high-side driver signal is then generated as a function of the combined response signal, such that the high-side driver signal substantially preserves the input information represented by the input signal, and such that the first exponential response and the second exponential response are substantially absent from the high-side driver signal.

CROSS-REFERENCES

This application is a continuation of U.S. patent application Ser. No.12/422,060, entitled “VOLTAGE LEVEL SHIFTER”, filed Apr. 10, 2009, whichclaims priority from U.S. Provisional Patent Application No. 61/044,113,filed Apr. 11, 2008, entitled “VOLTAGE LEVEL SHIFTER”, and from U.S.Provisional Patent Application No. 61/045,208, filed Apr. 15, 2008,entitled “VOLTAGE LEVEL SHIFTER FOR ARBITRARY INPUT SIGNALS”, the entiredisclosures of which are hereby incorporated by reference, as if setforth in full in this document, for all purposes.

BACKGROUND

The present invention relates to integrated circuits in general and, inparticular, to voltage level shifter circuits.

Many electronics applications use voltage level shifting and drivercomponents to handle high-side circuitry. Some of these applicationsprovide a high-side switch, where a load is switched at the high side(e.g., the voltage supply side) of a circuit. For example, when alow-power voltage source (e.g., a computer output, battery, etc.) isused to drive a potentially high-current load, it may be desirable toprovide a high-side switch that uses the low-power voltage source as acontrol signal to switch on another (e.g., higher-power) voltage sourceconnected to the load.

Other applications may use level shifting and driver components toconvert a direct current (“DC”) bus to an alternating current (“AC”)voltage for driving an AC system. For example, some uninterrupted powersupplies convert a DC bus voltage to a three-phase waveform for powerbackup functionality, some motor controllers convert one or more DC busvoltages into two- or three-phase control signals, and some solar cellsconvert generated DC voltages into AC voltages for standard householduses. Certain other applications convert an AC voltage to a DC bus,which may then be used to generate a switched voltage signal of one ormore frequencies. For example, when using a “Class D” audio amplifier todrive speakers, it may be desirable to increase the frequency of theamplifier, which may in turn decrease certain types of distortion (e.g.,by effectively increasing the sampling rate).

Certain limitations of many voltage level shifting circuits, however,may prevent the reliable operation of these types of applications athigh voltages and/or at high switching frequencies. One limitation isthat the types of components in the circuit may generate excessive heatat high voltages and/or high switching frequencies, which may causethermal run-away. Another limitation is that the configuration ofcomponents in the circuit may allow noise-induced cross conduction,which may short the DC bus voltage to ground. Either thermal run-away orshorting the DC bus to ground may, in some cases, cause permanent damageto the components and/or the packaging of the circuit.

As such, it may be desirable to provide voltage level shifting that mayoperate reliably and at low power, even at high voltages and highswitching frequencies.

SUMMARY

Among other things, methods, systems, and devices are described forproviding voltage level shifting, while avoiding excessive powerdissipation, cross-conduction, and/or other issues. Embodiments receivea two-level input signal representing input information, and effectivelygenerate two voltage responses as a function of the input signal. Thefirst voltage response includes a first exponential response definedsubstantially as a voltage across a parallel resistive-capacitive(“R-C”) network in response to a switched current. The second voltageresponse includes a second exponential response defined substantially asa voltage across the parallel R-C network in response to a switchedvoltage applied across an attenuator network including a secondcapacitive load coupled in series with the first network. A combinedresponse signal is generated substantially as a superposition of thefirst response signal and the second response signal. A high-side driversignal is then generated as a function of the combined response signal,such that the high-side driver signal substantially preserves the inputinformation represented by the input signal, and such that the firstexponential response and the second exponential response aresubstantially absent from the high-side driver signal.

In some embodiments, the first response is generated by: generating afirst switching voltage signal and a second switching voltage signal asa function of the input signal; generating a first switching currentsignal as a function of the first switching voltage signal; andgenerating a second switching current signal as a function of the secondswitching voltage signal. The second response is generated as a functionof the input signal by: when the second switching voltage signal isHIGH, building up a first charge reserve on a first precharging deviceand dumping at least a portion of a second charge reserve into thesecond current switching device; and, when the first switching voltagesignal is HIGH, building up the second charge reserve on a secondprecharging device and dumping at least a portion of the first chargereserve into the first current switching device. In certain embodiments,the combined response signal is generated by receiving the firstresponse signal and the second response signal differentially andisolating the first exponential response and the second exponentialresponse substantially to a common mode of the combined response signal.The high-side driver signal is generated as a function of the combinedresponse signal by rejecting the common mode of the combined responsesignal, such that the first exponential response and the secondexponential response are substantially absent from the high-side driversignal.

In one set of embodiments, a system is provided for voltage levelshifting. The system includes: an input module, operable to receive atwo-level input signal representing input information, and to generate afirst switching voltage signal and a second switching voltage signal asa function of the input signal; a current signal generator module,having: a first current switching device, operable to generate a firstswitching current signal as a function of the first switching voltagesignal; a second current switching device, operable to generate a secondswitching current signal as a function of the second switching voltagesignal; a first precharging device, coupled with the first currentswitching device and the second current switching device, and operableto build up a charge reserve when the second switching voltage signal isHIGH, and to dump at least a portion of the charge reserve into thefirst current switching device when the first switching voltage signalis HIGH; a second precharging device, coupled with the first currentswitching device and the second current switching device, and operableto build up a charge reserve when the first switching voltage signal isHIGH, and to dump at least a portion of the charge reserve into thesecond current switching device when the second switching voltage signalis HIGH; a voltage signal generator module, operable to generate a firstvoltage response as a function of the first switching current signal andto generate a second voltage response as a function of the secondswitching current signal; and a latching module, operable to generate atwo-level latched signal as a function of the first voltage response andthe second voltage response, such that the latched signal substantiallypreserves the input information represented by the input signal.

BRIEF DESCRIPTION OF THE DRAWINGS

A further understanding of the nature and advantages of the presentinvention may be realized by reference to the following drawings. In theappended figures, similar components or features may have the samereference label. Further, various components of the same type may bedistinguished by following the reference label by a second label thatdistinguishes among the similar components (e.g., a lower-casecharacter). If only the first reference label is used in thespecification, the description is applicable to any one of the similarcomponents having the same first reference label irrespective of thesecond reference label.

FIG. 1 shows a simplified block diagram of a system for using a voltagelevel shifter in an exemplary high-side switch configuration.

FIG. 2 shows a simplified block diagram of a system for using a voltagelevel shifter in an exemplary half-bridge configuration.

FIG. 3 shows a schematic view of an embodiment of a system for using avoltage level shifter in an exemplary half-bridge configuration, likethe one shown in FIG. 2.

FIG. 4 shows graphs of exemplary waveforms of signals read at certainpoints in the system shown in FIG. 3.

FIG. 5 shows a schematic view of an embodiment of a system for using avoltage level shifter for generating a combined voltage response in anexemplary half-bridge configuration, according to various embodiments ofthe invention.

FIG. 6 shows graphs of exemplary waveforms of signals read at certainpoints in the system shown in FIG. 5.

FIG. 7 shows a flow diagram of embodiments of voltage level shifting,according to various embodiments of the invention.

FIG. 8 shows a simplified block diagram of an illustrative voltage levelshifter configured to accept arbitrary input signals, according tovarious embodiments of the invention.

FIG. 9 shows a schematic view of an embodiment of an implementation of avoltage level shifter, like the one shown in FIG. 8, according tovarious embodiments of the invention.

FIG. 10 shows graphs of illustrative waveforms of signals read atcertain points in the voltage level shifter of FIG. 9.

FIG. 11 shows a flow diagram of exemplary methods for using a voltagelevel shifter with arbitrary input signals, according to embodiments ofthe invention.

DETAILED DESCRIPTION

Among other things, systems, devices, and methods are described forproviding voltage level shifting that may operate reliably and at lowpower, even at high voltages and/or high switching frequencies.

Many electronics applications use voltage level shifting, for example,with driver components to handle high-side circuitry. One set ofapplications provides a high-side switch, where a load is switched atthe high side (e.g., the voltage supply side) of a circuit. For example,when a low-power voltage source (e.g., a computer output, battery, etc.)is used to drive a potentially high-current load, it may be desirable toprovide a high-side switch that uses the low-power voltage source as acontrol signal to switch on another (e.g., higher-power) voltage sourceconnected to the load.

Another set of applications uses level shifting and driver components toconvert a direct current (“DC”) bus to an alternating current (“AC”)voltage for driving an AC system. For example, some uninterrupted powersupplies convert a DC bus voltage to a three-phase waveform for powerbackup functionality, some motor controllers convert one or more DC busvoltages into two- or three-phase control signals, and some solar cellsconvert generated DC voltages into AC voltages for standard householduses. Certain other applications convert an AC voltage to a DC bus,which may then be used to generate a switched voltage signal of one ormore frequencies. For example, when using a “Class D” audio amplifier todrive speakers, it may be desirable to increase the frequency of theamplifier, which may in turn decrease certain types of distortion (e.g.,by effectively increasing the sampling rate).

FIG. 1 shows a simplified block diagram of a system for using a voltagelevel shifter in an exemplary high-side switch configuration. The systemincludes a high-side switch 100 that receives a high-side controlvoltage 104 and drives a high-side switching device 150. The high-sideswitching device 150 is operable to switch an output voltage 160 acrossa load 165 between a bus voltage 102 and ground 108. The load may be anytype of resistive and/or reactive load, including, for example, a lamp,motor, heating coil, etc. The high-side switching device 150 may be anycompatible type of switching device, including a field effect transistor(“FET”), power metal-oxide FET (“power-MOSFET”), insulated gate bipolartransistor (“IGBT”), etc.

Some embodiments of the high-side switch 100 include a voltage levelshifter unit 110. One function of the voltage level shifter unit 110 maybe to provide and maintain the voltages and/or currents necessary todrive the high-side switching device 150 from the high-side controlvoltage 104. In one embodiment, the high-side switching device 150 is aFET, operable to be switched ON (i.e., to provide current to the load165) substantially in its linear region when the gate-to-source voltagefor the FET exceeds a threshold amount. Since the source of the FET istied to the output voltage 160, the gate voltage of the FET may have toexceed the output voltage 160 by the threshold amount.

It is worth noting that, while the high-side switching device 150 isOFF, the output voltage 160 may be pulled to ground 108 (e.g., by theload 165). In this state, a voltage may have to be applied to the gateof the FET that exceeds ground 108 by the threshold amount to turn ONthe high-side switching device 150. However, when the high-sideswitching device 150 is ON, the output voltage 160 may be pulled to thebus voltage 102 (e.g., 600 volts). In this state, the voltage applied tothe gate of the FET may now have to exceed the bus voltage 102 by thethreshold amount to keep the high-side switching device 150 ON.Embodiments include a high-side source 125, for example, for providingthe extra voltage necessary to pull up the gate of the high-sideswitching device 150. For example, the top of the high-side source 125may effectively provide a supply voltage for various components of thehigh-side switch 100 (e.g., with respect to the output voltage 160level).

Embodiments of the high-side switch 100 also include a high-side driverunit 120. The high-side driver unit 120 may be implemented in any usefulway, for example, including a transformer, discrete transistor,integrated circuit (“IC”), etc. The high-side driver unit 120 mayprovide a number of different functions, depending on the type ofhigh-side switching device 150 and other components being used and/orthe circuit configuration. Some embodiments of the high-side driver unit120 generate a high-side switching signal 145 for driving the high-sideswitching device 150. In other embodiments, the high-side driver unit120 controls high peak currents and heat dissipation generated by apower-MOSFET operating at high frequencies, as an isolation amplifier orshort-circuit protector for an IGBT, to provide a continuous gatecircuit for sustaining gate current in an IGBT, etc.

A number of components may be provided as part of, or in addition to,the voltage level shifter unit 110 to provide proper voltages andcurrents to the components in the system, like charge pumps, DC biasvoltage buses, etc. For example, in certain embodiments, input logic isprovided prior to the voltage level shifter unit 110 to interpret thehigh-side control voltage 104 and convert it into one or more signalsfor use by the voltage level shifter unit 110. In other embodiments, thevoltage level shifter unit 110 includes components for mitigating oreliminating potentially undesirable operation of the high-side switch100 For example, as explained in more detail below, using certainvoltage level shifter unit 110 topologies to generate the high-sideswitching signal 145 may create certain undesirable conditions. Onecondition may be excessive power dissipation due, for example, to highswitching currents used to support high switching frequencies. Anothercondition may be improper switching of certain components (e.g.,flip-flops) due, for example, to noise.

In some embodiments, the voltage level shifter unit 110 includes aprecharging unit 130 and/or a protection unit 140. Embodiments of theprecharging unit 130 are configured to reduce the amount of currentneeded to maintain a given switching frequency of the high-sideswitching signal 145 by precharging certain switching devices.Embodiments of the protection unit 140 are configured to minimizeimproper switching of the high-side switching signal 145 due to noise.Some embodiments of the protection unit 140 are further configured toallow faster recovery from improper switching to minimize undesirableeffects of the improper switching.

As shown, some embodiments of the voltage level shifter unit 110 includean input for receiving a low-side control voltage. When used in ahigh-side switch 100 configuration, the low-side control voltage may beprovided by a voltage source (e.g., a bias voltage tied to ground 108).It will be appreciated that other components may be needed for propercircuit functioning, even though they are not shown. For example,certain components may have source voltage specifications (e.g., certainlogic components may use 5-volt or 15-volt source voltages) or otherspecifications, as may be determined by certain design parameters or theuse of other components.

FIG. 2 shows a simplified block diagram of a system for using a voltagelevel shifter in an exemplary half-bridge configuration. The systemincludes a high-side switch 100 that receives a high-side controlvoltage 104 and drives a high-side switching device 150, and a low-sideswitch 200 that receives a low-side control voltage 204 and drives alow-side switching device 250. The high-side switching device 150 andthe low-side switching device 250 are configured as a half-bridge 270;the high-side switching device 150 is tied between a bus voltage 102 andan output voltage 160 (e.g., an output voltage bus), and the low-sideswitching device 250 is tied between the output voltage 160 and ground108. In this configuration, the half-bridge 270 may be operable toswitch the output voltage 160 between the bus voltage 102 and ground108.

In some embodiments, the high-side switch 100 is configured to operatelike the high-side switch 100 of FIG. 1. Embodiments of the high-sideswitch 100 include a voltage level shifter unit 110 in communicationwith a high-side driver unit 120. In certain embodiments, the voltagelevel shifter unit 110 includes a precharging unit 130 and/or aprotection unit 140, as described with reference to FIG. 1. Thehigh-side driver unit 120 is configured to generate a high-sideswitching signal 145 for driving the high-side switching device 150.

In the half-bridge 270 configuration of FIG. 2, the low-side controlvoltage 204 may include a control voltage signal, rather than a biasvoltage or other type of source (e.g., as may be the case in FIG. 1).The low-side control voltage 204 may be received by a low-side driverunit 220, configured to generate a low-side switching signal 245 fordriving the low-side switching device 250. Certain embodiments of thelow-side switch 200 further include a low-side source 225 for providingan appropriate source voltage to the low-side driver unit 220. Becausethe high-side control voltage 104 controls the high-side switchingdevice 150 (e.g., by driving the high-side switch 100 to generate ahigh-side switching signal 145) and the low-side control voltage 204controls the low-side switching device 250 (e.g., by driving thelow-side switch 200 to generate a low-side switching signal 245), it maybe preferable to configure the high-side control voltage 104 and thelow-side control voltage 204 such that only one of the high-sideswitching device 150 or the low-side switching device 250 may be ON(e.g., conducting) at any given time.

If both the high-side switching device 150 and the low-side switchingdevice 250 are ON at the same time, the bus voltage 102 may be shortedto ground 108. This may cause the circuit to malfunction, and may evencause permanent damage to one or more components. It will be appreciatedthat improper design of various components, and/or certain componentcharacteristics, may cause the high-side switching device 150 and thelow-side switching device 250 to be ON at the same time. For example,excessive propagation delay may adversely affect the timing between thehigh-side control voltage 104 and the low-side control voltage 204, ortheir propagation to their respective switching devices. In anotherexample, noise and/or other artifacts in the system may cause prematureswitching of certain components, or other issues, which may result inthe high-side switching device 150 and the low-side switching device 250being ON at the same time. In some embodiments, the low-side controlvoltage 204 is the output of a controller unit (not shown), operable,for example, to limit propagation delay, control cross-conduction, etc.

In certain embodiments, both the high-side switching device 150 and thelow-side switching device 250 are the same type of device (e.g., bothare NMOS devices). In these embodiments, preventing the devices frombeing ON at the same time may involve preventing the high-side switchingsignal 145 and the low-side switching signal 245 from being HIGH at thesame time. As such, in certain embodiments, the low-side control voltage204 is an inverted version of the high-side control voltage 104 (e.g.,the high-side control voltage 104 is passed through an inverter logicunit to generate the low-side control voltage 204).

In other embodiments, the high-side switching device 150 and thelow-side switching device 250 are different types of devices. Forexample, as shown in FIG. 2, the high-side switching device 150 may be aPMOS device and the low-side switching device 250 may be an NMOS device.In these embodiments, preventing the devices from being ON at the sametime may involve maintaining the high-side switching signal 145 and thelow-side switching signal 245 in the same state. For example, when boththe high-side switching signal 145 and the low-side switching signal 245are HIGH, the high-side switching device 150 may be OFF and the low-sideswitching device 250 may be ON. As such, in certain embodiments, thelow-side control voltage 204 and the high-side control voltage 104 maybe synchronized, tied together, inverted and re-inverted, etc.

It is worth noting that using a PMOS device for the high-side switchingdevice 150 (as in FIG. 2), as opposed to using an NMOS device for thehigh-side switching device 150 (as in FIG. 1), may require otheradjustments to circuit topologies. For example, in the high-side switch100 of FIG. 1, the high-side source 125 may provide a supply voltage forvarious components of the high-side switch 100 with respect to theoutput voltage 160 level. As shown in FIG. 2, however, the high-sidesource 125 effectively pulls down a reference level for components ofthe high-side switch 100 with respect to the DC bus level 102. Forexample, rather than using the output voltage 160 as a reference leveland using the high-side source 125 to provide a higher source voltagelevel for the high-side switch 100, the high-side switch 100 componentsare not connected to the output voltage 160 and are connected insteaddirectly to the DC bus voltage 102. Functionality relating to thetopology of FIG. 2 is described further with reference to FIG. 3.

FIG. 3 shows a schematic view of an embodiment of a system for using avoltage level shifter in an exemplary half-bridge configuration, likethe one shown in FIG. 2. FIG. 4 shows graphs of exemplary waveforms ofsignals read at certain points in the system shown in FIG. 3. For addedclarity, FIGS. 3 and 4 will be discussed in parallel.

The system 300 includes a high-side switch 100 that receives a high-sidecontrol voltage 104 and generates a high-side switching signal 145 fordriving a high-side switching device 150 (e.g., a first power-MOSFET),and a low-side switch 200 that receives a low-side control voltage 204and generates a low-side switching signal 245 for driving a low-sideswitching device 250 (e.g., a second power-MOSFET). The high-sideswitching device 150 and the low-side switching device 250 areconfigured as a half-bridge 270; the high-side switching device 150 istied between a bus voltage 102 and an output voltage 160 (e.g., anoutput voltage bus), and the low-side switching device 250 is tiedbetween the output voltage 160 and ground 108. In this configuration,the half-bridge 270 may be operable to switch the output voltage 160between the bus voltage 102 and ground 108.

It will be appreciated by those of skill in the art that the waveformsillustrated herein (e.g., in FIG. 4) may be presented in ideal orsimplified forms for added clarity. For example, where it does notmaterially add to the disclosure, the waveforms may be illustratedwithout delay, slope, slew rates, ringing, noise, etc. As such, thesimplified nature of the illustrative waveforms should not be construedas limiting the scope of the invention in any way.

The low-side control voltage 204 is shown in the first graph 402 of FIG.4, as a square wave, going from zero volts to a supply voltage(“V_(CC)”). The low-side control voltage 204 is passed through alow-side switch 200, including a low-side driver 220 energized by alow-side driver source 225 (e.g., a 15-volt DC). The output of thelow-side driver 220 is used to switch the gate voltage of the low-sideswitching device 250. This low-side gate voltage signal may besubstantially equivalent to the low-side control voltage 204, with somepropagation delay, as shown in the second graph 404. It will beappreciated that, based on properties of the low-side driver source 225and other components, the low-side gate voltage signal may differ fromthe low-side control voltage 204 in amplitude or other parameters.

The high-side control voltage 104 may be received by an input logicblock 305, which includes a number of logic units. In some embodiments,the high-side control voltage 104 is a square wave, going from zerovolts to a supply voltage (“V_(CC)”), as shown in the third graph 406 ofFIG. 4. As shown, the input logic block 305 includes inverter blocks303, delay blocks 307 and AND logic blocks 309. The input logic block305 may convert the high-side control voltage 104 into three additionalcontrol signals: an inverted high-side control voltage, a delayedhigh-side control voltage, and an inverted delayed high-side controlvoltage. The inverted high-side control voltage may be substantially aninverted version of the high-side control voltage 104; the delayedhigh-side control voltage may be substantially a delayed version of thehigh-side control voltage 104 (e.g., as shown in the fourth graph 408 ofFIG. 4); and the inverted delayed high-side control voltage may besubstantially an inverted version of the delayed high-side controlvoltage.

In the embodiment shown in FIG. 3, the inverted high-side controlvoltage is generated by passing the high-side control voltage 104through a first inverter block 303-1. The delayed high-side controlvoltage is generated by passing the high-side control voltage 104through the delay block 307. The inverted delayed high-side controlvoltage is generated by passing the delayed high-side control voltagethrough a second inverter block 303-2. The high-side control voltage 104and the inverted delayed high-side control voltage are passed to a firstAND logic block 309-1 to generate a first current switching signal. Theinverted high-side control voltage and the delayed high-side controlvoltage are passed to a second AND logic block 309-2 to generate asecond current switching signal.

In some embodiments, the voltage level shifter unit 110 includes a firsttransistor 312-1 and a second transistor 312-2. In one embodiment, thetwo transistors are low current, high voltage NMOS devices, both capableof withstanding the full bus voltage 102 (e.g., 600V between the drainand source of the transistor). The gate of the first transistor 312-1 isdriven with respect to its source (at ground 108) by the first currentswitching signal. The gate of the second transistor 312-2 is driven withrespect to its source (at ground 108) by the second current switchingsignal.

The gate voltages of the first transistor 312-1 and the secondtransistor 312-2 (e.g., the first current switching signal and thesecond current switching signal) are shown in the fifth graph 410 andsixth graph 412 of FIG. 4, respectively. It is worth noting that thegate voltages of the first transistor 312-1 and the second transistor312-2 are substantially pulse signals, each having a pulse widthaffected by the amount of delay between the high-side control voltage104 and the delayed high-side control voltage. In one embodiment, thepulses are narrow pulses, each having a pulse width of approximatelyfifty nanoseconds. When either the first transistor 312-1 or the secondtransistor 312-2 is driven by the fifty-nanosecond pulse, it may conductapproximately fifty milliamps during the time the pulse is in its HIGHstate, as shown in the seventh graph 414 and the eighth graph 416 ofFIG. 4, respectively.

It will be appreciated that there are many ways to generate pulse inputsfor turning ON the two transistors, the first transistor 312-1 and thesecond transistor 312-2. In certain embodiments, however, it may beimportant to ensure that the current switching signals are not HIGH atthe same time, such that the first transistor 312-1 and the secondtransistor 312-2 are not ON at the same time. For example, turning thefirst transistor 312-1 and the second transistor 312-2 ON at the sametime may cause SET and RESET inputs of a flip-flop (e.g., 316) to behigh at the same time, which may cause undesirable results, like causingthe high-side switching device 150 to switch ON while the low-sideswitching device 250 is ON. It will be further appreciated that theseand other undesirable results may be caused by artifacts of the circuitdesign, like dV/dt transitions, propagation delays, noise,cross-conduction, etc.

In one embodiment, when the high-side control voltage 104 goes HIGH, thegate of the first transistor 312-1 is driven to +15 volts (e.g., HIGH)for fifty nanoseconds, thereby conducting fifty milliamps. Thisfifty-milliamp current is converted to a negative going voltage across afirst resistor 313-1, which is clamped by a first zener diode 315-1.This clamped voltage drives the SET-bar input of a set-reset flip-flop316. The Q-bar output of the set-reset flip-flop 316 may then go LOW.When the high-side control voltage 104 goes LOW, the gate of the secondtransistor 312-2 is driven to +15 volts (e.g., HIGH) for fiftynanoseconds, thereby conducting fifty milliamps. This fifty-milliampcurrent is converted to a second negative going voltage across a secondresistor 313-2, which is clamped by a second zener diode 315-2. Thissecond clamped voltage drives the RESET-bar input of the set-resetflip-flop 316. The Q-bar output of the set-reset flip-flop 316 may thengo HIGH.

The Q-bar output waveform of the set-reset flip-flop 316 is shown in theninth graph 418 of FIG. 4 as a square wave that substantially followsthe high-side control voltage 104 (e.g., or an inverted version of thehigh-side control voltage 104) after some propagation delay. The Q-baroutput signal may pass through a high-side driver 120, driven by ahigh-side source 125, configured to generate a high-side switchingsignal 145 for use in driving the gate voltage of the high-sideswitching device 150 (e.g., turning the high-side switching device 150ON or OFF). An embodiment of the high-side switching signal 145 is shownin the tenth graph 420 of FIG. 4 as a square wave that substantiallyfollows the Q-bar output waveform after some propagation delay.

When the high-side switching signal 145 goes LOW, the high-sideswitching device 150 may turn ON (e.g., the high-side switching device150 is shown as a PMOS device). In the ON state, the high-side switchingdevice 150 may act substantially like a closed circuit, conductingcurrent and pulling the output voltage 160 to the bus voltage 102. Whenthe high-side gate driving voltage goes HIGH, the high-side switchingdevice 150 may turn OFF. In the OFF state, the high-side switchingdevice 150 may act substantially like an open circuit, preventingcurrent from flowing. When the low-side gate voltage signal goes HIGH(as shown in the second graph 404 of FIG. 4), the low-side switchingdevice 250 may turn ON (e.g., the low-side switching device 250 is shownas an NMOS device). In the ON state, the low-side switching device 250may act substantially like a closed circuit, conducting current andpulling the output voltage 160 to ground 108. It will be appreciatedthat, depending on the load attached to the output voltage 160,parasitic capacitance of the devices, and other factors, the outputvoltage 160 may essentially remain at or near the bus voltage 102 untilthe voltage is sinked by the low-side switching device 250 or some otherdevice. An embodiment of the output voltage 160 waveform is shown in theeleventh graph 422 of FIG. 4.

The twelfth graph 424 and the thirteenth graph 426 of FIG. 4 illustratethe drain-source voltages across the first transistor 312-1 and thesecond transistor 312-2, of the voltage level shifter unit 310,respectively. By examining the twelfth graph 424 in the context of theseventh graph 414, it may be seen that the first transistor 312-1 maytypically be conducting current only when there is little or nodrain-source voltage across the first transistor 312-1. As such, thepulse power whenever the first transistor 312-1 turns on may berelatively small (e.g., and may be ignored for many practical purposes).

However, by examining the thirteenth graph 426 in the context of theeighth graph 416, it may be seen that the second transistor 312-2 maytypically be conducting current while the drain-source voltage acrossthe second transistor 312-2 is approximately the full bus voltage 102(or even higher). In some typical applications, the bus voltage 102 maybe approximately 600 volts, and the pulse width of the gate voltage forthe second transistor 312-2 may be approximately 50 ns, generating apulse power of approximately thirty watts (i.e., 50 mA*600V). Manyswitching applications may desire to operate at switching frequencies ofone Megahertz or higher. At a switching frequency one Megahertz, theaverage power dissipation of the second transistor 312-2 may becalculated as 1.5 watts (i.e., 30 W*50 ns*1 MHz). Because manyintegrated circuits may be rated to handle around 0.5 watts, this levelof power dissipation may require special packaging technologies tomaintain safe device operating temperatures without permanent devicedamage or destruction, when operating at these voltages and/orfrequencies.

Particularly, components of the voltage level shifter unit 110 maymanifest capacitive properties (e.g., stray capacitance). For example,each transistor 312 may manifest a so-called Miller capacitance, and theinputs to the set-reset flip-flop may manifest stray capacitance. As thetransistors switch, voltage transitions in their respective currentpaths may be slowed by the capacitive effects on the resistor-capacitor(“R-C”) time constant of the voltage level shifter unit 110.

This effect may be illustrated by analyzing components of the circuit ina simplified form as a switched current source (e.g., the transistors312 driven by the current switching signals) driving a capacitive load,C_(L) (e.g., the stray capacitances), in parallel with a resistive load,R_(L) (e.g., resistors 313). The switched current source provides acurrent step signal that transitions from zero to a positive currentvalue, I_(L) at an initial time (t=0). A voltage step response for thevoltage across the parallel network (e.g., the voltage across theresistive load and the capacitive load) may be calculated as:

$\begin{matrix}{{V_{LI}(t)} = {I_{L}*R_{L}*{\left( {1 - {\mathbb{e}}^{{- t}*{(\frac{1}{R_{L}*C_{L}})}}} \right).}}} & \;\end{matrix}$

The voltage step response illustrates that the R-C time constant causesthe voltage transition to occur over a period of time. It will beappreciated that, at least for this reason, higher switching currentsmay typically be used to ensure adequate bandwidth for supporting higherswitching frequencies. These high currents may cause or exacerbate thepower dissipation issues discussed above.

In some embodiments, the voltage level shifter unit 110 includes aprecharging unit 130, configured to allow operation of the high-sideswitch 100 at high switching frequencies, while using lower currents.Reducing the amount of current needed may reduce the power dissipationof the circuit. This may, for example, allow the circuit to be used withstandard IC processes (e.g., typically lower cost components andmanufacturing processes). In some embodiments, the precharging unit 130includes a first capacitor 314-1 and a second capacitor 314-2. The firstcapacitor 314-1 is connected between the SET-bar input of the set-resetflip-flop 316 (the drain of the first transistor 312-1) and the gate ofthe second transistor 312-2. The second capacitor 314-2 is connectedbetween the RESET-bar input of the set-reset flip-flop 316 (the drain ofthe second transistor 312-2) and the gate of the first transistor 312-1.

In this configuration, the switched transistor 312 topology may beanalyzed substantially as a switched voltage source with respect to thecapacitors 314. For example, turning the first transistor 312-1 ON maycause the second capacitor 314-2 to charge. When the first transistor312-1 is turned OFF, and the second transistor 312-2 is turned ON,stored charge in the second capacitor 314-2 may be dumped into thesecond transistor 312-2. In this way, the second capacitor 314-2 mayeffectively precharge the second transistor 312-2, which may offsetstray capacitive effects. At the same time, turning the secondtransistor 312-2 ON may cause the first capacitor 314-1 to charge. Whenthe second transistor 312-1 is turned OFF again, and the firsttransistor 312-1 is turned ON again, stored charge in the firstcapacitor 314-1 may be dumped into the first transistor 312-1.

These effects may be illustrated by analyzing components of the circuitin a simplified form, including the precharging unit 130, as a switchedvoltage source (e.g., the transistors 312 driven by the currentswitching signals) driving a first capacitive load, C₁ (e.g., thecapacitors 314 in the precharging unit 130), in series with a parallelnetwork having a second capacitive load, C₂ (e.g., the straycapacitances), in parallel with a resistive load, R_(L) (e.g., resistors313). The switched voltage source provides a voltage step signal thattransitions from zero to some positive voltage value, V_(L), at aninitial time (t=0). The voltage step response of the voltage across theparallel network with the second capacitive load and the resistive loadmay be calculated as:

${V_{LV}(t)} = {{V_{IN}\left( {t = 0^{+}} \right)}*\left( \frac{C\; 1}{{C\; 1} + {C\; 2}} \right)*{\left( {\mathbb{e}}^{{- t}*{(\frac{1}{R_{L}*{({{C\; 1} + {C\; 2}})}}}} \right).}}$

Again, the voltage step response illustrates that the R-C time constantcauses the voltage transition to occur over a period of time. Notably,the current switching effects may cause the voltage step response toexponentially transition from zero volts to a steady state level over atime defined by the R-C time constant. However, the voltage switchingeffects may cause the voltage step response to exponentially transitionfrom the steady state level to zero volts over substantially the sametime defined by the R-C time constant. The topology shown in FIG. 3,including the precharging unit 130, may be configured to allow thevoltage switching and current switching effects to effectively besuperimposed (e.g., setting C₁+C₂ equal to C_(L), and operating thecomponents within their linear ranges). Superimposing the effects mayillustrate that the precharging unit 130 can be used to offset straycapacitive effects of components of the voltage level shifter unit 110.For example, the superimposed effects may be calculated as follows:

$\begin{matrix}{{V_{L}(t)} = {{V_{LI}(t)} + {V_{LV}(t)}}} \\{= {\left\lbrack {{I_{IN}\left( {t = 0^{+}} \right)}*R_{L}*\left( {1 - {\mathbb{e}}^{{- t}*{(\frac{1}{R_{L}*C_{L}})}}} \right)} \right\rbrack +}} \\{\left\lbrack {{V_{IN}\left( {t = 0^{+}} \right)}*\left( \frac{C\; 1}{C_{L}} \right)*\left( {\mathbb{e}}^{{- t}*{(\frac{1}{R_{L}*C_{L}})}} \right)} \right\rbrack.}\end{matrix}$

It will be appreciated that, according to the combined load voltageresponse equation just after the initial time (at t=0⁺), the combinedload voltage response may be calculated as:

$\begin{matrix}{{V_{L}\left( {t = 0^{+}} \right)} = {\left\lbrack {{I_{IN}\left( {t = 0^{+}} \right)}*R_{L}*\left( {1 - 1} \right)} \right\rbrack + \left\lbrack {{V_{IN}\left( {t = 0^{+}} \right)}*\left( \frac{C\; 1}{C_{L}} \right)*(1)} \right\rbrack}} \\{= {{V_{IN}\left( {t = 0^{+}} \right)}*{\left( \frac{C\; 1}{C_{L}} \right).}}}\end{matrix}$

Further, according to the combined load voltage response equation atsteady state (e.g., t=∞), the combined load voltage response may becalculated as:

$\begin{matrix}{{V_{L}\left( {t = \infty} \right)} = {\left\lbrack {{I_{IN}\left( {t = 0^{+}} \right)}*R_{L}*\left( {1 - 0} \right)} \right\rbrack + \left\lbrack {{V_{IN}\left( {t = 0^{+}} \right)}*\left( \frac{C\; 1}{C_{L}} \right)*(0)} \right\rbrack}} \\{= {{I_{IN}\left( {t = 0^{+}} \right)}*{R_{L}.}}}\end{matrix}$

Embodiments are configured to set the current through the transistor 312paths (e.g., by sizing a resistor and a voltage source accordingly) suchthat:

${{I_{IN}\left( {t = 0^{+}} \right)}*R_{L}} = {{V_{IN}\left( {t = 0^{+}} \right)}*{\left( \frac{C\; 1}{C_{L}} \right).}}$

The result may then be calculated as:

${V_{L}\left( {t = 0^{-}} \right)} = {{0\mspace{14mu}{and}\mspace{14mu}{V_{L}\left( {t = 0^{+}} \right)}} = {{V_{IN}\left( {t \geq 0^{+}} \right)}*{\left( \frac{C\; 1}{C_{L}} \right).}}}$

These equations illustrate that, by adding the voltage responses fromthe current switching circuit and the voltage switching circuit, acombined load voltage response may be generated that manifestsessentially a step response attenuated by the ratio of C1/C_(L) (e.g.,the exponential terms of the individual responses may be effectivelyeliminated by adding the responses in this way). This may provide asleast two features.

First, because the exponential effects of the individual responses maylimit the bandwidth of the voltage level shifter unit 110. Including theprecharging unit 130 of FIG. 3, however, may mitigate the exponentialeffects, which may allow operation of the voltage level shifter unit 110at lower currents for a given switching frequency. As described above,high currents and/or switching frequencies may cause certain devices(e.g., the second transistor 312-2) to generate excessive self-heating,which may cause thermal run-away and possible permanent damage. Allowingoperation at lower currents may, in effect, reduce or eliminate thesepower dissipation issues for certain applications.

Second, it may be desirable for the high-side switching signal 145 tomanifest substantially a step response. Because the voltage response ofthe voltage level shifter unit 110 may include exponential terms (e.g.,because of the exponential effects seen when the precharging unit 130 isnot present), embodiments of high-side switches 100 use digital latchingtechniques (e.g., the set-reset flip-flop 316) to generate the stepresponse. Use of digital latching devices, however, may cause certainundesirable results.

One such undesirable result is that the devices may be prone tonoise-induced cross conduction, which may allow both the high-sideswitching device 150 and the low-side switching device 250 to be ON atthe same time, potentially shorting the bus voltage 102 to ground 108.This cross conduction may, for example, be induced by dV/dt transientnoise that exceeds some threshold value (e.g., typically around ±50V/ns) or by propagation delays. Further, because of the latching, it maybe difficult or impossible to recover from the improper switchingconfigurations (e.g., it may be necessary to wait for another switchingcycle to effectively reset the set-reset flip-flop 316). Another suchundesirable result is that the latching devices may manifestunpredictable conditions at startup. For example, without additionalcircuitry, the set-reset flip-flop 316 may start up in a condition thatallows the high-side switching device 150 and the low-side switchingdevice 250 to be ON at the same time. As such, some embodiments includean under-voltage lock-out unit 322 to cause the high-side switch 100 tostart up in a predetermined, desirable condition. The extra circuitrymay add complexity and/or expense to the fabrication of the circuit insome cases.

For at least these reasons, it may be desirable to avoid use of digitallatching techniques and their associated devices. Notably, thesuperimposing effects of the precharging unit 130 may cause the voltageresponse of the voltage level shifter unit 110 to manifest a stepresponse, as discussed above. This may indicate that an appropriatecircuit configuration with an appropriately set bias current for thetransistor 312 paths may be used to generate a high-side switchingsignal 145 that also manifests a step response, without using digitallatching techniques, like the set-reset flip-flop 316.

FIG. 5 shows a schematic view of an embodiment of a system for using avoltage level shifter for generating a combined voltage response in anexemplary half-bridge configuration, according to various embodiments ofthe invention. As discussed below, the system 100 is optimized toexploit the combined load voltage response effects of using prechargingunits (e.g., the precharging unit 130 of FIG. 3). For example, thesystem 100 is shown to avoid use of digital latching techniques or theirassociated devices (e.g., there is no set-reset flip-flop 316 orunder-voltage lock-out unit 322, as in the system 300 of FIG. 3). FIG. 6shows graphs of exemplary waveforms of signals read at certain points inthe system shown in FIG. 5. For added clarity, FIGS. 5 and 6 will bediscussed in parallel.

The system 500 includes a high-side switch 100 that receives a high-sidecontrol voltage 104 and generates a high-side switching signal 145 fordriving a high-side switching device 150 (e.g., a first power-MOSFET),and a low-side switch 200 that receives a low-side control voltage 204and generates a low-side switching signal 245 for driving a low-sideswitching device 250 (e.g., a second power-MOSFET). The high-sideswitching device 150 and the low-side switching device 250 areconfigured as a half-bridge 270; the high-side switching device 150 istied between a bus voltage 102 and an output voltage 160 (e.g., anoutput voltage bus), and the low-side switching device 250 is tiedbetween the output voltage 160 and ground 108. In this configuration,the half-bridge 270 may be operable to switch the output voltage 160between the bus voltage 102 and ground 108.

In some embodiments, the low-side control voltage 204 is a square wave,going from zero volts to a supply voltage (“V_(CC)”). The low-sidecontrol voltage 204 is passed through a low-side switch 200, including alow-side driver 220 energized by a low-side driver source 225 (e.g., a15-volt DC source). The output of the low-side driver 220 is used togenerate the low-side switching signal 245 for switching the gatevoltage of the low-side switching device 250. In certain embodiments,the low-side switching signal 245 is substantially equivalent to thelow-side control voltage 204, with some propagation delay. In otherembodiments, the low-side switching signal may differ from the low-sidecontrol voltage 204 in amplitude or other parameters.

The high-side switch 100 includes a voltage level shifter unit 110 and ahigh-side gate driver 120. It will be appreciated by those of skill inthe art that the configuration of the voltage level shifter unit 110 mayessentially include functionality of two switching voltage sources andtwo switching current sources, with their effects superimposed, asdescribed above. In some embodiments, the two switching voltage sourcesare provided by receiving a high-side control voltage 104 (e.g., asquare wave) and passing the high-side control voltage 104 through aninverter 510 or other logic to generate an inverted high-side controlvoltage 504. Embodiments of the high-side control voltage 104 and theinverted high-side control voltage 504 are shown in the first graph 602and the second graph 604 of FIG. 6, respectively. As shown, the controlvoltages may be square waves transitioning between zero volts and aninput HIGH voltage level, “V_(IN).”

The high-side control voltage 104 and the inverted high-side controlvoltage 504 may then be used as two complementary switching voltagesources. In some embodiments, the two switching current sources areprovided by using a pair of complementary transistors 312 connected to acurrent source 516. When each transistor 312 turns ON in turn, it mayconduct current according to the current source 516. If the gates of thepair of transistors 312 are driven by complementary square wave voltagesignals (e.g., the high-side control voltage 104 and the invertedhigh-side control voltage 504), the transistors 312 may generateessentially complementary square wave current signals. Embodiments ofcurrent through the first transistor 312-1 and the current through thesecond transistor 312-2 are shown in the third graph 606 and the fourthgraph 608 of FIG. 6, respectively. As shown, the current signals may besquare waves transitioning between zero amps and the bias current levelprovided by the current source 516.

In the embodiments shown in FIG. 5, the high-side control voltage 104and the inverted high-side control voltage 504 are used as two switchingvoltage sources. Two transistors, the first transistor 312-1 and thesecond transistor 312-2 are provided, one side of each being connectedto a current source 516 set to draw an amount of current (e.g., a biascurrent). The gate of the first transistor 312-1 is driven by thehigh-side control voltage 104 and the gate of the second transistor312-2 is driven by the inverted high-side control voltage 504. In thisconfiguration, the current waveforms of the first transistor 312-1 andthe second transistor 312-2 may substantially follow the voltagewaveforms of the high-side control voltage 104 and the invertedhigh-side control voltage 504, respectively.

It will be appreciated that, while this and other embodiments aredescribed with reference to square wave control signals (e.g., thehigh-side control voltage 104), any arbitrary waveform and/or anyarbitrary duty cycle is possible according to the invention. In oneembodiment, the high-side control voltage 104 is a square pulse of10%/90% duty cycle. The result may be a difference in direct current(“DC”) offset from a 50% duty cycle square wave, which may vary as afunction of the time constants (e.g., the resistor-capacitor timeconstant with respect to the repetition frequency). For example, if thetime constant were ten nanoseconds for a pulse repetition frequency of100 Kilohertz (i.e., a ten microsecond period), the offset may benegligible; but if the same ten-nanosecond time constant were applied toa ten Megahertz signal (i.e., a 100 nanosecond period), a DC offset mayresult between the pulses.

The voltage level shifter unit 110 may further include a first networkof passive devices, including capacitor 314-1, capacitor 512-1, resistor514-3, and resistor 514-1, and a second network of passive devices,including capacitor 314-2, capacitor 512-2, resistor 514-2, and resistor514-1. In some embodiments, capacitor 314-1 and capacitor 314-2 areconfigured as a precharging unit 130; and, in certain embodiment,capacitor 512-1 and capacitor 512-2 are configured as attenuators. Thehigh-side control voltage 104 and the current waveform generated by thesecond transistor 312-2 may be used to control the first network ofpassive devices, thereby generating a first combined response. Theinverted high-side control voltage 504 and the current waveformgenerated by the first transistor 312-1 may be used to control thesecond network of passive devices, thereby generating a second combinedresponse.

In one embodiment, the high-side control voltage 104 drives the gate ofthe first transistor 312-1 and the inverted high-side control voltage504 drives the gate of the second transistor 312-2. A high goinghigh-side control voltage 104 steers a tail current provided by thecurrent source 516 (e.g., fifty micro-amps) through the first transistor312-1, thereby driving a drain load resistor, resistor 514-2. The samecurrent may flow through resistor 514-1, reaching a high-side sourcevoltage terminal sitting at a voltage level generated by a high-sidesource 125. The high-side source 125 may generate a voltage level of“V_(CC),” and may be connected between the high-side source voltageterminal and an output voltage 160 level, such that the high-side sourcevoltage terminal is maintained at a level of the output voltage 160 plusV_(CC). A low going high-side control voltage 104 (i.e., a high-goinginverted high-side control voltage 504) steers the current provided bythe current source 516 through the second transistor 312-2, therebydriving a second drain load resistor, resistor 514-3. Again, the samecurrent may flow through resistor 514-1, reaching the high-side sourcevoltage terminal.

As the sourced current will either flow through resistor 514-2 when thefirst transistor 312-1 is turned on or through resistor 514-3 when thesecond transistor 312-2 is turned on, there may be a substantiallyconstant current (e.g., substantially the full bias current provided bythe current source 516) flowing through resistor 514-1. The value ofresistor 514-1 may be chosen so that approximately half of the V_(CC)voltage generated by the high-side source 125 will be dropped acrossresistor 514-1. The turning on of the first transistor 312-1 may cause anegatively going transient across resistor 514-2, and the turning off ofthe second transistor 312-2 may cause a positively going transientacross resistor 514-2. These transients may be capacitively reinforcedby capacitor 314-1 and capacitor 314-2. A negatively going transient maybe reinforced by capacitor 314-2 as a result of the inverted high-sidecontrol voltage 504, and a positively going transient may be reinforcedby capacitor 314-1 as a result of the non-inverted high-side controlvoltage 104.

For example, the voltage across resistor 514-2 may be calculated toproduce a waveform like the one shown in the fifth graph 620 of FIG. 6.It will be appreciated that the voltage across resistor 514-3 may becalculated to produce a waveform that is essentially the complement ofthe one shown in the sixth graph 630 of FIG. 6. While the waveforms mayinclude exponential terms, appropriately designing and implementing thevoltage level shifter unit 110 circuit may allow the exponential termsof the voltage switching circuitry and the current switching circuitryto be isolated to the common mode of the first combined response (e.g.,graph 620) and the second combined response (e.g., graph 630).

Embodiments are configured so that the common mode exponential terms maybe effectively rejected by using the signals differentially. Thedifferential response 545 (e.g., the voltage seen differentially at theinputs to a hysteresis comparator 540) may look like the waveform shownin the seventh graph 640 of FIG. 6. As shown in the seventh graph 640,the hysteresis comparator 540 effectively sees a step response at itsinput with substantially no exponential terms. This may be a result ofthe common mode rejection capabilities of the hysteresis comparator 540.For example, the hysteresis comparator 540 may have a highcharacteristic common mode rejection ratio, allowing the device torecognize small changes in the desirable portion of the differentialinput voltage, while rejecting relatively large fluctuations in thecommon mode of the differential input voltage. In this way, thehysteresis comparator 540 may be configured as a protection unit 140.

In some embodiments, the protection unit 140 can be further construed asincluding capacitor 512-1 and capacitor 512-2. For example, actualfluctuations in source voltage 102 levels may be much larger than thecommon mode rejection capabilities of the hysteresis comparator 540.However, capacitor 512-1 and capacitor 512-2 may be configured (e.g.,their values may be selected) to attenuate the effects seen at thedifferent inputs of the hysteresis comparator 540, for example, by afactor of 100. It is worth noting that the differential response 545 isessentially a bi-polar step response (e.g., going from the negativevoltage drop across resistor 514-2 to the positive voltage drop acrossresistor 514-3) that substantially follows the combined load voltageresponse equation derived above.

In some embodiments, the differential response 545 is used todifferentially drive the hysteresis comparator 540. The hysteresiscomparator 540 may be operable to compare the voltages at its twoinputs. When its positive input voltage exceeds its negative inputvoltage by some positive threshold amount, the hysteresis comparator 540may output a logical HIGH voltage; and when its negative input voltageexceeds its positive input voltage by some negative threshold amount,the hysteresis comparator 540 may output a logical LOW voltage. It isworth noting that the positive threshold value and the negativethreshold value may be set at any practical and useful voltage fordifferent reasons. For example, it may be desirable to set either orboth of the positive threshold value and the negative threshold value toavoid undesirable transitions due to noise in the system (e.g., dV/dtnoise).

The output of the hysteresis comparator 540 may be communicated to thehigh-side driver 120 to generate a high-side switching signal 145.Embodiments of the high-side driver 120 are connected between the outputvoltage 160 and the high-side source voltage terminal. The high-sidedriver 120 may be configured to provide an appropriate level forswitching the high-side switching device 150 as a function of the outputof the comparator. An embodiment of the high-side switching signal 145is shown in the eighth graph 650 of FIG. 6. Embodiments of the high-sideswitching signal 145 may be complementary signals to embodiments of thelow-side switching signal 245.

The high-side switching signal 145 may be used to drive the gate of thehigh-side switching device 150 and the low-side switching signal 245 maybe used to drive the gate of the low-side switching device 250. In thisconfiguration, the high-side switching signal 145 and the low-sideswitching signal 245 may essentially control the half-bridge 270 toswitch the output voltage 160 between the bus voltage 102 and ground108. An embodiment of the output voltage 160 may look like the waveformshown in the ninth graph 660 of FIG. 6.

It will be appreciated that certain component values (or ratios) may beselected to provide certain results. For example, the value of thedifferential response may be set (e.g., as a result of capacitor 314-1and capacitor 314-2) to be substantially equal to the current providedby the current source 516 (“I_(BIAS)”) times the value of the resistor514-2. This may result in little or no delay between the high-sidecontrol voltage 104 and the differential response (e.g., due to thefirst transistor 312-1, the second transistor 312-2, and theirassociated parasitic substrate capacitances). As another example,component ratios may be set such that:

${\frac{V_{DC\_ BUS}}{\frac{1}{2}V_{CC}} = {\frac{C_{314 - 1}}{C_{512 - 1}} = {\frac{C_{512 - 2}}{C_{314 - 2}} = {\frac{V_{CC}}{I_{BIAS}*R_{514 - 2}} \leq K}}}},$where K is a constant value.

In some embodiments, delay between the high-side control voltage 104 andthe output voltage 160 (input-to-output delay) may be primarily due toresponse delay of the comparator. As is known in the art, comparatordelay may decrease exponentially as its input overdrive is increased.Setting V_(CC) to fifteen volts and K to 100, for example, thedifferential response 145 (i.e., the differential input to thehysteresis comparator 540) may be calculated as 300 millivolts. Further,using a bus voltage of 600 volts and setting K to 100 may cause aninduced common mode response seen at the input of the comparator to becalculated as approximately six volts (i.e., V_(BUS)/100=600V/100), andthe lowest input common mode voltage to be calculated as 1.6 volts(i.e., (V_(CC)/2)−(V_(BUS)/101)=7.5−5.9). These values may be kept wellwithin a rated input range of the comparator.

It will be further appreciated that the power dissipation of the voltagelevel shifter unit 110 may essentially be calculated as the time eitherthe first transistor 312-1 or the second transistor 312-2 is on and isexperiencing the full bus voltage 102 (e.g., 600V). A worst case may bewhen the output voltage 160 is pulled to the bus voltage 102 for most ofeach cycle of the high-side control voltage 104. In this case, the powerdissipation may essentially be calculated as thirty milliwatts (i.e., 50μA*600V). It is worth noting that thirty milliwatts may be well withinmany standard IC package technologies for self-dissipation of heat withsimple convection cooling techniques, as known in the art.

It will now be appreciated by those of skill in the art that using avoltage level shifter unit, like the voltage level shifter unit 110shown in FIG. 5, may avoid some of the undesirable results inherent withdigital latching techniques. In one example, the low power dissipationmay avoid the thermal runaway experienced by some digitally latchedvoltage level shifters used at high voltages and/or frequencies. Inanother example, because the power dissipation is apparently independentof (or constant with) frequency, using the device at high switchingfrequencies may not generate excessive heat. In yet another example,because the device does not use a digitally latched technique, it may beself-correcting after experiencing any temporary noise transients beyondits rated dV/dt. As such, the device may be able to accept dV/dttransitions of plus or minus fifty volts-per-nanosecond, or higher,without error. In still another example, the configuration of thecircuit may eliminate the need for an under-voltage lock-out circuit,which may reduce the cost and/or complexity of the circuitimplementation.

FIG. 7 shows a flow diagram of more specific embodiments of voltagelevel shifting, according to various embodiments of the invention. Themethod 700 may begin by receiving a high-side control voltage. At block710, the high-side control voltage is used to generate two switchingvoltage signals and two switching current signals. The first switchingvoltage signal may be tied to the first switching current signal, andthe second switching voltage signal may be tied to the second switchingcurrent signal. The two switching voltage signals may be configured sothat only one of the first or second switching voltage signals is ON atany time.

At block 720, the two switching voltage signals and the two switchingcurrent signals are passed through two circuit networks to generate twocombined response signals. Each of the circuit networks may be operableto combine the functionality of a current switching circuit and avoltage switching circuit, such that each combined response signal iseffectively a combination of a response signal from a current switchingcircuit and a response signal from a voltage switching circuit. The twocombined response signals are used in block 730 to differentially drivea comparator and generate a comparator output. The comparator output ispassed through a high-side gate driver at block 740 to generate ahigh-side gate driver signal. The high-side gate driver signal is usedat block 750 to switch a high-side switching device. In someembodiments, the high-side switching device is configured for use as ahigh-side switch. In other embodiments, the high-side switching deviceis configured for use as part of a half bridge.

Voltage Level Shifter Embodiments for Arbitrary Input Signals

The embodiments described above with reference to FIGS. 1-7 areoptimized for handling two-level (e.g., digital) signals. For example,various embodiments include logic units, switching signals, and otherdigital types of implementations. It may be desirable, in someapplications to level shift arbitrary (e.g., analog) input signals.Embodiments described with reference to FIGS. 8-11 provide voltage levelshifting functionality for arbitrary input signals. In some embodiments,the level-shifted output may accurately represent the arbitrary inputsignal information even in the context of an unstable reference.

Many electronics applications use voltage level shifting as part ofdetection and/or isolation circuitry. Some of these applications providecircuitry that detects or receives signals from one system with onereference voltage, and level shifts the signal to another system withanother reference voltage. For example, it may be desirable to use asmall-signal input voltage to provide information to a relativelyhigh-voltage system. To ensure that the information from the smallsignal voltage may be used by the high-voltage system, it may benecessary to level shift the voltage. Level shifting the voltage mayhelp, for example, to reject large-signal common-mode voltages that mayinterfere with the accurate detection of the small-signal information.Further, level shifting the voltage may allow the small-signal systemthat generated the small-signal input voltage to be electricallyisolated from the high-voltage system.

In one illustrative case, it is desirable to detect current passing intothe motor of an electric vehicle. A current sensor may be placed inseries with the motor input, such that a voltage signal is generated,the voltage signal being proportional to the input current to the motor.The full range of the generated voltage signal may typically be on theorder of only a few volts. The generated voltage signal may be passed toa signal processing system configured to adjust certain vehicleparameters depending on the input current to the motor. The signalprocessing system may operate in an electrical environment where itsreference voltage fluctuates by hundreds of volts. As such, thegenerated voltage signal may be essentially in the noise of the signalprocessing system, and the large voltage fluctuations of the signalprocessing system may adversely affect the motor input system if thesystems are not isolated from each other. For these and/or otherreasons, it may be desirable to voltage shift the generated voltagesignal, such that the voltage shifted signal essentially rides on top ofthe fluctuating reference voltage of the signal processing system whileremaining electrically isolated from the system that created thegenerated voltage signal.

FIG. 8 shows a simplified block diagram of an illustrative voltage levelshifter configured to accept arbitrary input signals, according tovarious embodiments of the invention. The voltage level shifter 800includes a voltage-to-current converter unit 810, a current-to-voltageconverter unit 820, and a gain stage 830. The voltage level shifter 800receives two complementary inputs, a voltage input signal 802, and aninverted voltage input signal 806, both with respect to a firstreference voltage 808 (e.g., ground). In some embodiments, the invertedvoltage input signal 806 is generated by transforming the voltage inputsignal 802. In one embodiment, the voltage input signal 802 is passedthrough an inverting amplifier 804 to generate the inverted voltageinput signal 806. Other ways of generating complementary input signalsare known in the art.

In some embodiments, the voltage input signal 802 and the invertedvoltage input signal 806 are received by the voltage-to-currentconverter unit 810. The voltage-to-current converter unit 810 maytransform the received voltage signals 802 and 806 into at least onecurrent signal, representing the information from the received voltagesignals 802 and 806. It will be appreciated that the transformation maycause the generated current signal(s) to differ from the receivedvoltage signals 802 and 806, for example, in phase and/or amplitude.

The generated current signal(s) may then be received by thecurrent-to-voltage converter unit 820. The current-to-voltage converterunit 820 may transform the generated current signal(s) into at least onegenerated voltage signal. The generated voltage signal(s) may representthe information from the current signal(s). As with thevoltage-to-current converter unit 810, the transformation by thecurrent-to-voltage converter unit 820 may cause the generated voltagesignal(s) to differ from the generated current signal(s), for example,in phase and/or amplitude. In some embodiments, the transformation bythe current-to-voltage converter unit 820 may substantially be theinverse of the transformation by the voltage-to-current converter unit810.

The generated voltage signal(s) may be used to drive the gain stage 830of the voltage level shifter 800 (e.g., differentially). In certainembodiments, the gain stage 830 includes a differential amplifier, whilein other embodiments, the gain stage 830 includes an analog-to-digitalconverter. Other types of compatible gain stage components are known inthe art. The output of the gain stage 830 may represent the differencebetween the voltages seen at its input terminals (e.g., the differencebetween two generated voltage signals). For example, the gain stage 830may be driven in such a way as to effectively recreate the voltage inputsignal 802. The output of the gain stage 830 may then be used as avoltage output signal 860 of the voltage level shifter 800.

In certain embodiments, the gain stage 830 may provide additionalfunctionality. One additional function of the gain stage 830 may be toaffect the gain of its output (e.g., to amplify the voltage outputsignal 860). Another additional function of the gain stage 830 may be tohelp electrically isolate the voltage output signal 860 from the sourceof the voltage input signal 802 and/or other components. A thirdadditional function of the gain stage 830 may be to provide impedancematching between the voltage level shifter 800 (or the source of thevoltage input signal 802) and any other systems that may be electricallyconnected with the gain stage 830.

In some embodiments, the gain stage 830 is tied between a bias voltageand a reference voltage 850, via a bias voltage source 840. In certainembodiments, the reference voltage 850 is tied to a hard point (e.g., aground reference), while in other embodiments, the reference voltage 850floats. Because the gain stage 830 is referenced to the referencevoltage 850, the voltage output signal 860 may float on the referencevoltage 850. So long as the gain stage 830 has sufficient common-moderejection capabilities, this may allow the gain stage 830 to effectivelyreject fluctuations in the reference voltage 850. This, in turn, mayallow information from the voltage output signal 860 to be used withoutbeing affected by fluctuations in the reference voltage 850 inundesirable ways.

FIG. 9 shows a schematic view of an embodiment of an implementation ofthe voltage level shifter 800 shown in FIG. 8, according to variousembodiments of the invention. FIG. 10 shows graphs of illustrativewaveforms of signals read at certain points in the circuit 900 of FIG.9. For added clarity, FIGS. 9 and 10 will be discussed in parallel.

The voltage level shifter 800 may receive two complementary inputvoltages at a voltage-to-current converter unit 810. In someembodiments, the voltage-to-current converter unit 810 includes currentgain components operable to receive voltage inputs and generateproportional current outputs. In one embodiment, the voltage-to-currentconverter unit 810 includes a first transistor 912-1, a secondtransistor 912-2, a first current transforming resistor 914-1, a secondcurrent transforming resistor 914-2, and a current source 916. Thecurrent source 916 may be configured to maintain a substantiallyconstant bias current (“I_(BIAS)”), and may be tied to a voltagereference (e.g., ground 808).

In some embodiments, the input voltages are provided by receiving avoltage input signal 802 (e.g., an arbitrary, analog waveform), andpassing the voltage input signal 802 through an inverting amplifier 804to generate an inverted voltage input signal 806. The voltage inputsignal 802 and the inverted voltage input signal 806 may then be used ascomplementary input voltages. An illustrative embodiment of a voltageinput signal 802 is shown in the first graph 1002 of FIG. 10, as anarbitrary, analog signal. An illustrative embodiment of a complementaryvoltage input signal 906 is shown in the second graph 1004 of FIG. 10,as an arbitrary, analog signal that is the complement of the signalshown in the first graph 1002.

In some embodiments, the voltage input signal 802 drives the gate of afirst transistor 912-1, and the inverted voltage input signal 806 drivesthe gate of a second transistor 912-2. The first transistor 912-1 is inseries with a first current transforming resistor 914-1, and the secondtransistor 912-2 is in series with a second current transformingresistor 914-2. In certain embodiments, the values of the currenttransforming resistors 914 are selected to provide high conductance withrespect to the mutual conductance of the transistors 912. The currenttransforming resistors 914 may be tied to the current source 916. Inthis way, the gain of the transistors 912 may be substantially greaterthan the conductance of the current transforming resistors 914, whichmay allow the current through the transistors 912 to be substantiallyproportional to the voltages at their gates.

It will be appreciated that, in this configuration, the transistors 912may be operable to provide complementary current signals thateffectively represent the complementary voltage signals provided by thevoltage input signal 802 and the inverted voltage input signal 806.Illustrative embodiments of a first generated current signal flowingthrough the first transistor 912-1 and a second generated current signalflowing through the second transistor 912-2 are shown in the third graph1006 and the fourth graph 1008 of FIG. 10, respectively. It is worthnoting that the first generated current signal and the second generatedcurrent signal straddle a current of ½*I_(BIAS), half of the currentprovided by the current source 916. As such, the addition of the firstgenerated current signal to its complementary second generated currentsignal may result in a substantially constant current of ½*I_(BIAS).

Embodiments of the voltage level shifter 800 may receive the firstgenerated current signal and the second generated current signal at acurrent-to-voltage converter unit 820. The current-to-voltage converterunit 820 may further receive the voltage input signal 802 and theinverted voltage input signal 806. The current-to-voltage converter unit820 may include a first network of passive devices, including capacitor918-1, capacitor 918-3, resistor 914-5, and resistor 914-3, and a secondnetwork of passive devices, including capacitor 918-2, capacitor 918-4,resistor 914-4, and resistor 914-3. The voltage input signal 802 and thesecond generated current signal may be used to control the first networkof passive devices, thereby generating a first generated voltage signal.The inverted voltage input signal 806 and the first generated currentsignal may be used to control the second network of passive devices,thereby generating a second generated voltage signal.

In one embodiment, the voltage input signal 802 drives the gate of thefirst transistor 912-1 and the inverted voltage input signal 806 drivesthe gate of the second transistor 912-2. As the voltage input signal 802increases, the first transistor 912-1 may allow more current to flow(i.e., the first generated current signal amplitude increases), therebycausing more current to flow through a first drain load resistor,resistor 914-4. At the same time, the increasing voltage input signal802 may generate a decreasing inverted voltage input signal 806 (sincethe two voltages are complementary), which may decrease the current flowthrough the second transistor 912-2 and through a second drain loadresistor, resistor 914-5. Because both resistor 914-4 and resistor 914-5are in series with resistor 914-3, and both are in series with thecurrent source 916 of the voltage-to-current converter unit 810, thecurrent through resistor 914-3 may remain substantially constant.Resistor 914-3, capacitor 918-1, and capacitor 918-2 are furtherconnected to a bias voltage source 840. The bias voltage source may beconfigured to generate a bias voltage 970 that is a given level above areference voltage 850. As such, the voltage drop across resistor 914-3may remain substantially constant, as determined by the current source916 and the bias voltage source 840. For example, the value of resistor914-3 may be chosen so that approximately half of the bias voltage(generated by the bias voltage source 840) will be dropped acrossresistor 914-3.

It is worth noting that the changes in the first generated currentsignal and the second generated current signal may cause positive andnegative voltage transients across resistor 914-4 and resistor 914-5.The positive transients may be capacitively reinforced by capacitor918-3 as a result of the voltage input signal 802, and the negativetransients may be reinforced by capacitor 918-4 as a result of theinverted voltage input signal 806. For example, waveform distortioncreated by capacitor 918-1 and capacitor 918-2, those working withrespect to load resistors resistor 914-4 and resistor 914-5, iscancelled by capacitor 918-3 and capacitor 918-4 (e.g., as a“feed-forward” circuit). This cancellation may be assisted by selectingvalues of various components such that, for example, the value ofresistor 914-4 equals the value of resistor 914-5, the value of resistor914-1 equals the value of resistor 914-2, and

${\frac{C_{918 - 3}}{C_{918 - 3} + C_{918 - 1}} = \frac{R_{914 - 4}}{R_{914 - 4} + R_{914 - 1}}},$where, for example, “C₉₁₈₋₃” represents the value of capacitor 918-3.

It will be appreciated that the cancellation may not occur for thecommon mode voltage developed across resistor 914-3 with respect to thereference voltage 850. For example, this may be because the referencevoltage 850 effectively acts as a common mode noise generator when thereference voltage 850 is floating. However, this may not adverselyimpact the output of the circuit, where the common mode voltagedeveloped across resistor 914-3 remains less than the input common moderange of the gain stage 830 (e.g., within the common mode rejectioncapabilities of the gain stage 830). It is worth noting that the gainstage 830 may be connected between the bias voltage 970 and thereference voltage 850.

Illustrative embodiments of a first generated voltage signal across thefirst drain resistor, resistor 914-4, and a second generated voltagesignal across the second drain resistor, resistor 914-5, are shown inthe fifth graph 1010 and the sixth graph 1012 of FIG. 10, respectively.It is worth noting that the first generated voltage signal straddles avoltage calculated as the value of resistor 914-4 times half of the biascurrent (i.e., R₉₁₄₋₄*½*I_(BIAS)), and the second generated voltagesignal straddles a voltage calculated as the value of resistor 914-5times half of the bias current (i.e., R₉₁₄₋₅*½*I_(BIAS)). As such, ifresistor 914-4 and resistor 914-5 are selected to be of equal value andthe second generated voltage signal is subtracted from its complementaryfirst generated voltage signal, a differential voltage response 932 maybe calculated as the value of resistor 914-3 times the bias current(i.e., R₉₁₄₋₃*I_(BIAS)).

In some embodiments, the differential voltage 932 may be used to drive again stage 830. The gain stage 830 may include a differential amplifier,an analog to digital converter, or any other compatible component. Thegain stage 830 may be used for any of various functions, including togenerate an output voltage 860 from the differential voltage 932, toaffect the gain (e.g., to amplify) the output voltage 860, to impedancematch the output voltage 860, etc.

In certain embodiments, the gain stage 830 is tied between the biasvoltage and the reference voltage 850, via the bias voltage source 840.In certain embodiments, the reference voltage 850 is tied to a hardpoint (e.g., a ground reference), while in other embodiments, thereference voltage 850 floats. Because the gain stage 830 is referencedto the reference voltage 850, the voltage output signal 860 may float onthe reference voltage 850. So long as the gain stage 830 has sufficientcommon-mode rejection capabilities, this may allow the gain stage 830 toeffectively reject fluctuations in the reference voltage 850. This, inturn, may allow information from the voltage output signal 860 to beused without being affected by fluctuations in the reference voltage 850in undesirable ways. The output voltage, then, may look like thewaveform shown in the seventh graph 1014 of FIG. 10. As shown, theoutput voltage waveform shown in graph 1014 may retain substantially allthe information of the input voltage waveform shown in graph 1002.Notably, however, the waveforms may ride on different reference levels.For example, while the input voltage waveform may ride on a relativelystable chassis ground, the output level may ride on a widely fluctuatingfloating ground reference.

FIG. 11 shows a flow diagram of exemplary methods for using a voltagelevel shifter, according to embodiments of the invention. The method1100 begins by receiving an arbitrary input voltage signal at block1110. At block 1120, the arbitrary input voltage signal is converted(e.g., transformed) into at least one generated current signal thatrepresents the information from the arbitrary input voltage signal. Atblock 1130, the at least one generated current signal is converted intoat least one generated voltage signal. The at least one generatedvoltage signal is used to differentially drive a gain stage and generatea level-shifted voltage signal at block 1140. The level shifted voltagemay be output at block 1150 as an output voltage.

It should be noted that the methods, systems, and devices discussedabove are intended merely to be examples. It must be stressed thatvarious embodiments may omit, substitute, or add various procedures orcomponents as appropriate. For instance, it should be appreciated that,in alternative embodiments, the methods may be performed in an orderdifferent from that described, and that various steps may be added,omitted, or combined. Also, features described with respect to certainembodiments may be combined in various other embodiments. Differentaspects and elements of the embodiments may be combined in a similarmanner. Also, it should be emphasized that technology evolves and, thus,many of the elements are examples and should not be interpreted to limitthe scope of the invention.

It should also be appreciated that the following systems, methods, andsoftware may individually or collectively be components of a largersystem, wherein other procedures may take precedence over or otherwisemodify their application. Also, a number of steps may be requiredbefore, after, or concurrently with the following embodiments.

Specific details are given in the description to provide a thoroughunderstanding of the embodiments. However, it will be understood by oneof ordinary skill in the art that the embodiments may be practicedwithout these specific details. For example, well-known circuits,processes, algorithms, structures, waveforms, and techniques have beenshown without unnecessary detail in order to avoid obscuring theembodiments. It will be further understood by one of ordinary skill inthe art that the embodiments may be practiced with substantialequivalents or other configurations. For example, circuits describedwith reference to N-channel transistors may also be implemented withP-channel devices, using modifications that are well known to those ofskill in the art.

Also, it is noted that the embodiments may be described as a processwhich is depicted as a flow diagram or block diagram. Although each maydescribe the operations as a sequential process, many of the operationscan be performed in parallel or concurrently. In addition, the order ofthe operations may be rearranged. A process may have additional stepsnot included in the figure.

Having described several embodiments, it will be recognized by those ofskill in the art that various modifications, alternative constructions,and equivalents may be used without departing from the spirit of theinvention. For example, the above elements may merely be a component ofa larger system, wherein other rules may take precedence over orotherwise modify the application of the invention. Also, a number ofsteps may be undertaken before, during, or after the above elements areconsidered.

Accordingly, the above description should not be taken as limiting thescope of the invention, as described in the following claims:

1. A voltage level shifting system comprising: a conversion subsystem,configured to convert an input voltage signal into complementary inputsignals and to use the complementary input signals to drive activedevices, each active device manifesting a dynamic current resulting fromparasitic capacitance associated with the active device; a compensationsubsystem, coupled with the conversion subsystem, and configured togenerate compensation currents to compensate for the dynamic currentsassociated with the active devices and to generate compensatedcomplementary signals corresponding to the complementary input signals;and an output subsystem, coupled with the compensation subsystem, andconfigured to generate an output voltage signal from the compensatedcomplementary signals, such that the output voltage signal is a levelshifted version of the input voltage signal.
 2. The voltage levelshifting system of claim 1, wherein the conversion subsystem isconfigured to convert the input voltage signal into two complementaryinput signals, each used to drive one of two active devices, each activedevice configured to generate a complementary current signal thatincludes a signal current corresponding to its respective complementaryinput signal and the dynamic current resulting from its parasiticcapacitance.
 3. The voltage level shifting system of claim 1, whereinthe compensation subsystem is configured to generate the compensationcurrents using the complementary input signals.
 4. The voltage levelshifting system of claim 1, wherein the compensation subsystem comprisesa first half-circuit and a second half-circuit in communication with afirst active device and a second active device of the conversionsubsystem, the first half-circuit having a resistive network incommunication with the first active device and a capacitive network incommunication with the second active device, and the second half-circuithaving a resistive network in communication with the second activedevice and a capacitive network in communication with the first activedevice, at least a portion of each capacitive network being designed tocompensate for the parasitic capacitance associated with a respectiveone of the active devices.
 5. The voltage level shifting system of claim1, wherein the compensation subsystem comprises: a first capacitivedevice configured to manifest a capacitance substantially equivalent tothe parasitic capacitance of a second of the active devices, the secondof the active devices being driven by a complement of the input voltagesignal and having a drain terminal coupled with a second active devicenode, the first capacitive device being coupled between the secondactive device node and the input voltage signal so as to generate afirst compensation current to compensate for the dynamic current of thesecond of the active devices; and a second capacitive device configuredto manifest a capacitance substantially equivalent to the parasiticcapacitance of a first of the active devices, the first of the activedevices being driven by the input voltage signal and having a drainterminal coupled with a first active device node, the second capacitivedevice being coupled between the first active device node and thecomplement of the input voltage signal so as to generate a secondcompensation current to compensate for the dynamic current of the firstof the active devices.
 6. The voltage level shifting system of claim 1,wherein the output system comprises a gain stage configured to receivethe compensated complementary signals differentially so as to generatethe output voltage signal according to a differential level between thecompensated complementary signals.
 7. The voltage level shifting systemof claim 1, wherein the conversion subsystem comprises: a first activedevice being a transistor driven by a first complementary input signalbeing the input voltage signal, a source terminal of the first activedevice being coupled with a first resistor that is further coupled witha current source, and a drain terminal of the first active device beingcoupled with a first active device node; and a second active devicebeing a transistor driven by a second complementary input signal beinggenerated by passing the input voltage signal through an inverter, asource terminal of the second active device being coupled with a secondresistor that is further coupled with the current source, and a drainterminal of the second active device being coupled with a second activedevice node.
 8. The voltage level shifting system of claim 7, whereinthe compensation subsystem comprises: a first capacitor coupled betweenthe first active device node and the second complementary input signal;a second capacitor coupled between the first active device node and abus voltage; a third capacitor coupled between the second active devicenode and the first complementary input signal; a fourth capacitorcoupled between the second active device node and the bus voltage; athird resistor coupled between the first active device node and a fourthresistor that is further coupled with the bus voltage; and a fifthresistor coupled between the second active device node and the fourthresistor.
 9. The voltage level shifting system of claim 8, wherein theoutput subsystem comprises: an operational amplifier having a firstdifferential input node coupled with the second complementary inputsignal and a second differential input node coupled with the firstcomplementary input signal, and configured to generate the outputvoltage signal according to a differential level between thedifferential input nodes, and the operational amplifier being tiedbetween the bus voltage and a reference level.
 10. The voltage levelshifting system of claim 1, wherein the input voltage signal comprises anon-periodic analog signal.
 11. The voltage level shifting system ofclaim 1, wherein the output voltage signal represents information fromthe input voltage signal while having a different phase or amplitude.12. The voltage level shifting system of claim 1, further comprising: apackaging configured to house at least the compensation subsystem.
 13. Amethod comprising: driving active devices using complementary inputsignals, each active device manifesting a complementary current signalcorresponding to a respective complementary input signal and a dynamiccurrent resulting from parasitic capacitance associated with the activedevice; generating compensation currents to compensate for the dynamiccurrents associated with the active devices; generating compensatedcomplementary signals corresponding to the complementary input signalsusing the complementary current signals compensated by the compensationcurrents; and generating an output voltage signal from the compensatedcomplementary signals, such that the output voltage signal is a levelshifted version of the input voltage signal.
 14. The method of claim 13,further comprising: converting an input voltage signal into thecomplementary input signals.
 15. The method of claim 13, whereingenerating compensation currents to compensate for the dynamic currentsassociated with the active devices comprises: generating a firstcompensation current to compensate for the dynamic current of a secondof the active devices using a first capacitive device configured tomanifest a capacitance substantially equivalent to the parasiticcapacitance of the second of the active devices, the second of theactive devices being driven by a complement of the input voltage signaland having a drain terminal coupled with a second active device node,the first capacitive device being coupled between the second activedevice node and the input voltage signal; and generating a secondcompensation current to compensate for the dynamic current of a first ofthe active devices using a second capacitive device configured tomanifest a capacitance substantially equivalent to the parasiticcapacitance of the first of the active devices, the first of the activedevices being driven by the input voltage signal and having a drainterminal coupled with a first active device node, the second capacitivedevice being coupled between the first active device node and thecomplement of the input voltage signal.
 16. The method of claim 13,wherein generating an output voltage signal from the compensatedcomplementary signals, such that the output voltage signal is a levelshifted version of the input voltage signal comprises: receiving thecompensated complementary signals differentially at a gain stage so asto generate the output voltage signal according to a differential levelbetween the compensated complementary signals.
 17. A method comprising:providing a compensation subsystem configured to generate compensationcurrents to compensate for dynamic currents associated with activedevices and to generate compensated complementary signals correspondingto the complementary input signals, the complementary input signalsbeing generated according to an input voltage signal and configured todrive the active devices so that each generates a complementary currentsignal corresponding to a respective one of the complementary inputsignals and its dynamic current resulting from an associated parasiticcapacitance, each compensated complementary signal being generatedaccording to a respective complementary current signal and compensatedby a respective compensation current, such that the compensatedcomplementary signals are usable differentially to generate an outputvoltage signal that is a level shifted version of the input voltagesignal.
 18. A system comprising: means for driving active devices usingcomplementary input signals, each active device manifesting acomplementary current signal corresponding to a respective complementaryinput signal and a dynamic current resulting from parasitic capacitanceassociated with the active device; means for generating compensationcurrents to compensate for the dynamic currents associated with theactive devices; means for generating compensated complementary signalscorresponding to the complementary input signals using the complementarycurrent signals compensated by the compensation currents; and means forgenerating an output voltage signal from the compensated complementarysignals, such that the output voltage signal is a level shifted versionof the input voltage signal.
 19. The system of claim 18, furthercomprising: means for converting an input voltage signal into thecomplementary input signals.
 20. The system of claim 18, wherein themeans for generating the output voltage signal from the compensatedcomplementary signals, such that the output voltage signal is a levelshifted version of the input voltage signal comprises: means forreceiving the compensated complementary signals differentially at a gainstage so as to generate the output voltage signal according to adifferential level between the compensated complementary signals.